mum stable gain (MSG) of 9 dB at 76 GHz. Here, fmax was determined from the maximum available gain curve instead of by -6 dB extrapolation of the unilateral power gain curve. This result shows that the new HEMT structure effectively reduces the parasitic effects. Using the 0.15-μm gate HEMT, we developed an MMIC process for low-cost millimeter wave applications. Similar to the conventional MMICs, MIM capacitors, epitaxial resistors, and air bridge wiring are employed for passive elements. The conventional MMIC uses micro-strips as transmission lines, which need a ground plane on the backside of the chip and via holes to connect the backside ground plane to the face-side of the chip. The thickness of the chip must be precisely controlled to maintain the impedance of the transmission line. This requires a wafer thinning process and wafer penetrating process for the fabrication. The controllability of the process to determine wafer thickness and via size poses a difficult problem to produce much higher frequency circuits with the micro-strip structure. A co-planar waveguide (CPW) is often used in millimeter-wave systems and its structure is also convenient for flip-chip
assembling, since all electrodes in the circuit are placed only on one side of the chip. No wafer thinning or via penetrating process is needed for coplanar circuits. The chip can also be thick, thus retaining mechanical strength during the chip bonding process. Digital LSIs usually use a bump technology based on transferred solder for the interconnections of flip-chips. The MMIC for the W-band requires more precise positioning and size of interconnection electrodes to avoid degradation due to impedance mismatching than digital LSIs because of its short wavelength. In order to obtain good positioning with a small diameter interconnection, we developed a pillar technique that is a part of the fabrication process. After the metallizing process for the MMIC co-planar lines, pillar-shaped electrodes with 40-μm diameter gold are formed on the wafer by electroplating. Figure 3 shows an SEM photograph of the MMIC, showing the
The mixer has a singly balanced configuration. The RF and LO signals are fed into the gate of each HEMT through the 0-p hybrid circuit. The 0-p hybrid circuit consists of a quarter-wavelength transmission line and a branch line hybrid circuit. The mixer using the 0-p hybrid circuit exhibits good isolation between the RF and LO ports even if the unit mixers have large reflection coefficients. Also, the mixer reduces AM noise from the LO in cooperation with an external IF 180- degree hybrid circuit. The mixer has a conversion gain of -4 dB at a 76-GHz LO frequency for an LO power of -1 dBm. The 76-GHz SPDT switch consists of two parallel- resonated 160-μm HEMTs. The transmission line connected to the drain and source terminals of the HEMT is employed as a resonant inductor in parallel with the fringing capacitance. The obtained insertion loss and isolation at 76.5 GHz were 3.4 dB and -17 dB, respectively. The 38/76-GHz doubler chip consists of two single-ended 80-μm HEMT frequency doublers and a branch line hybrid circuit. This circuit con- figuration leads to a low input return and high isolation between the two output ports. Figure 11 shows the measured frequency response of the doubler. The doubler has a conversion gain of -3.7 dB for a 38.25-GHz input signal power of 3 dBm. In the 38-GHz voltage controlled oscillator, we adopted the reverse channel HEMT configuration to generate sufficient negative resistance and reduce the size of the source feedback circuit. The oscillator uses the transmission line resonator terminated by a 160-μm Schottky-barrier diode as a varactor. The 38-GHz buffer amplifier provides impedance isolation between the oscillator and the external load. The chip has an output power of -0.7 dBm at 39.6 GHz. Figure 12 shows an SEM photo of the flipchip MMIC, indicating the amplifier chip on the evaluation board.
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