PARA MAS INFORMACION ACERCA DE RF and Microwave Active Device Technologies
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By Edgar Alberto Servita 18.856.338
CAF
Mostrando entradas con la etiqueta 2010-1 CAF Edgar Alberto Servita. Mostrar todas las entradas
Mostrando entradas con la etiqueta 2010-1 CAF Edgar Alberto Servita. Mostrar todas las entradas
domingo, 30 de mayo de 2010
Semiconductor Diodes
Semiconductor Diodes
The Semiconductor Diode
The semiconductor diode is a device that will conduct current in one direction only. It is the electrical equivalent of a hydraulic check valve. The semiconductor diode has the following characteristics:
· A diode is a two-layer semiconductor consisting of an Anode comprised of P-Type semiconductor material and a Cathode which is made of N-Type semiconductor material.
· The P-Type material contains charge carriers which are of a positive polarity and are known as holes. In the
N-Type material the charge carriers are electrons which are negative in polarity.
· When a semiconductor diode is manufactured, the P-Type and N-Type materials are adjacent to one another creating a P-N Junction.
iasing
A bias refers to the application of an external voltage to a semiconductor. There are two ways a P-N junction can be biased.
· A forward bias results in current flow through the diode (diode conducts). To forward bias a diode, a positive voltage is applied to the Anode lead ( which connects to P-Type material) and the negative voltage is applied to the Cathode lead ( which connects to N-Type material).
· A reverse bias results in no current flow through the diode (diode blocks). A diode is reverse biased when the
Anode lead is made negative and the Cathode lead is made positive.
P-N Junction Characteristics
The P-N Junction region has three important characteristics:
1) The junction is region itself has no charge carriers and is known as a depletion region.
2) The junction (depletion) region has a physical thickness that varies with the applied voltage. A forward
bias decreases the thickness of the depletion region; a reverse bias increases the thickness of the depletion region.
3) There is a voltage, or potential hill, associated with the junction. Approximately 0.3 of a volt is required to forward bias a germanium diode; 0.5 to 0.7 of a volt is required to forward bias a silicon diode.
Silicon Diodes
Ratings
Three characteristics must be defined for proper application or replacement of a semiconductor diode:
Voltage Rating is the maximum voltage which the diode will block in the reverse-biased mode.
· This is expressed as the Peak-Reverse-Voltage (PRV) or Peak-Inverse-Voltage (PIV).
· It is important to remember that this is a peak value of voltage not the root-mean-square (RMS) value. As a
"Rule -of-Thumb, to provide a margin of safety, the PIV rating of a diode should be at least 3 times the RMS voltage of the circuit.
Current Rating is the maximum current the device can carry in the forward biased direction.
Package Configuration
· Small, low current diodes are available in an axial lead configuration. The band end is the cathode.
· High current diodes come in a press-fit, stud- mounted, or hockey puck package.
Stud mounted diodes are available in Standard Polarity (stud cathode) and Reverse Polarity (stud anode).
Thermal Limits
· It is essential that semiconductors operate within the device temperature ratings.
· Semiconductor charge carriers are released thermally as well as electrically. Heat-sinking may be required during soldering and when the device is in operation to prevent thermal damage.
· The forward resistance of a diode decreases with temperature; this results in an increase in current, which in turn produces more heat. As a result, thermal run-away can occur and destroy the semiconductor.
Unlike its predecessor, the Analog Ohmmeter, Digital Ohmmeters require
a special Diode Check Function because the current circulated by the normal Ohms Function of a digital meter is too low to adequately check a diode.
In the Diode Check Position, the reading given by a digital meter in the forward bias direction (meter positive to diode anode and meter negative to diode cathode) is actually the voltage required to overcome the internal diode junction potential. For a silicon diode this will be about 0.5 -
0.8 volt; a germanium diode will read slightly lower, about 0.3 - 0.5 volt.
Symbol Notation K (or C) = Cathode, A = Anode.
Diode Test Procedure
Caution: Ohms and Diode Check measurements can be made only on de-energized circuits! The Ohmmeter
battery provides power to make this measurement. You may need to remove the diode from the circuit to get a
reliable test. See Note below.
· Connect leads to meter as shown - Black COM, Red W .
· Select the (Diode Test) function.
· Connect the leads to the Diode-Under-Test as shown in the drawing above and verify the readings are correct for both a forward and reverse bias. (This is sometimes referred to as checking the front-to-back ratio.)
Note: Large Stud-Mounted Diodes are bolted to a heat sink and Hockey Puck Units are compressed between the heat sinks; removing them from the circuit can be time-consuming and may be unnecessary. In these situations, test the entire assembly first, then, if the assembly tests shorted, remove and test the diodes individually. Hockey
Puck Diodes must be compressed in a heat sink assembly or test fixture to be tested as they require compression to make-up the internal connections.
Rectification
· Rectification is the process of converting an Alternating Current (AC) to a Direct Current (DC).
· In the circuits below, the DC output voltage is defined as pulsating DC because it has the same waveform as one-half cycle of the applied alternating current. It is DC because it always has the same polarity with respect to zero volts. On single-phase rectifiers, the output DC voltage goes to zero after each rectified half cycle.
· To convert a pulsating DC to a pure DC, such as that produced by a battery or DC generator, the DC output voltage must be filtered.
· The diode symbol points in the direction of conventional current flow (positive to negative).
· To analyze the operation of a rectifier circuit supplied by an AC circuit, arbitrarily assign a polarity to the transformer winding and analyze the diode operation, then reverse the polarity assignment and again analyze the operation of the diode. When the anode of the diode is made positive with respect to the cathode the diode will conduct. When the anode of the diode is made negative with respect to the cathode the diode will block the flow of current.
· When the diode is conducting, current flows through the diode and the voltage drop across the diode is very small (typically 0.5 - 0.7 volts for a silicon diode). The current flow through the load resistor produces a voltage drop across the load resistor.
· When the diode is non-conducting, no current flows through the diode and the applied voltage appears across the diode. Because there is no current flow, there will be no voltage drop across the resistor.
Three-Phase Half-Wave Rectifier
On three-phase rectifiers, the pulsations do not return to zero as with a single phase rectifier. This reduces the amount of ripple and simplifies filtering.
A diode is forward biased when the anode is made more positive with respect to the cathode. Each of the diodes is forward biased when the voltage of the phase leading it becomes lower than the diode anode voltage and the diode is reverse biased when the voltage of the phase lagging it becomes higher that diode anode voltage.
Three-Phase Full-Wave Rectifier
Showing rectifier transformer delta secondary only. When the diodes are replaced with SCR's, the output voltage of the rectifier can be controlled by phase-firing of the SCR's. This arrangement is referred to as a six-pulse system.
Six-Phase Systems
Some special medium-voltage rectifier transformers have dual secondary windings - one delta, the other wye - which are 30 degrees out-of-phase. The phase-to-phase voltage of the wye matches the phase voltage of the delta. The outputs are individually rectified and the rectifiers are connected in series, resulting in a six-phase system with very low ripple, that has an output voltage which is double the voltage of the individual windings. The dashed line in the corner of the delta shows the phase shift between the two windings.
Metalorganic vapor phase epitaxy of high-quality GaAs0.5Sb0.5
Metalorganic vapor phase epitaxy of high-quality GaAs0.5Sb0.5
and its application to heterostructure bipolar transistors
We report the growth and characterization of high-quality InP/GaAs0.5Sb0.5 /InP heterostructures and their application to double-heterojunction bipolar transistors ~DHBT!. The GaAs0.5Sb0.5 layer quality was evaluated by high-resolution x-ray diffraction ~XRD!, low-temperature photoluminescence ~PL!, and atomic force microscopy ~AFM!. The observed 4.2 K PL linewidth was 7.7 meV and XRD rocking curves matched those of dynamical scattering simulations. In contrast to previously reported InP/GaAs0.5Sb0.5 /InP DHBTs, the present devices show nearly ideal base and collector currents, low turn-on and collector offset voltages, and a high current gain.
Self-aligned DHBTs exhibit a cutoff frequency over 75 GHz and common-emitter current gain greater than 100 at 300 K.
GaAs0.5Sb0.5 lattice matched to InP substrates is a promising base material for NpN double heterojunction bipolar
transistors1,2 ~DHBTs! because of the very favorable band alignment at the InP/GaAs0.5Sb0.5 heterojunction. Recent low temperature photoluminescence measurements indicate that the GaAs0.5Sb0.5 conduction band is about ;180 meV above that of InP.3 The valence band offset is about 760 meV between GaAsSb and InP,3 which is two times larger than that of InGaAs/InP ~378 meV!. The large valence band offset effectively blocks hole back injection into the emitter. This band lineup eliminates any possible collector current blocking effect at the base-collector junction,4 and electrons are thus injected ballistically from the GaAs0.5Sb0.5 base to the InP collector. Wide band gap InP collector HBTs can then be grown without compositional grading at the base/collector junction, thus greatly simplifying device design and fabrication.
GaAsSb layers have been grown by metalorganic vapor phase epitaxy ~MOVPE!,1,2,5–8 but only Bhat et al.1 and Mc-
Dermott et al.2 reported the implementation of HBT prototypes based on this material system. These seminal reports of GaAsSb/InP HBTs featured poor junction ideality factors ~1.26–2.4! and a low current gain of about 22,1,2 which was then attributed to acceptor-like deep levels in the GaAs0.5Sb0.5 base layer. In addition, there are few reports concerning the optical properties of GaAsSb grown by MOVPE. In this letter, we report the growth and characterization of high-quality GaAs0.5Sb0.5 layers. Our InP/GaAsSb/ InP DHBTs show significantly improved direct current ~DC! and radio frequency ~RF! device characteristics. Epitaxial layers were grown in a horizontal reactor with a H2 total flow of 6 SLM at a reactor pressure of 100 Torr. Trimethylindium ~TMIn!, triethylgallium ~TEGa!, tertiarybutylarsine ~TBAs!, tertiarybutylphosphine ~TBP!, and trimethylantimony ~TMSb! were used as precursors. N-type doping was accomplished with H2S ~200 ppm in H2) and p type by CCl4 ~500 ppm in H2). The susceptor was kept at 560 °C by a resistance heater. The growth rates were ;1.0mm/h for InP, and ;1.3mm/h for undoped GaAs0.5Sb0.5 . The substrates were ~001! exactly oriented InP ''Epi-Ready'' Sumitomo wafers. At the GaAsSb/InP interface, both group III and V elements change. In order to facilitate the gasswitching scheme, a thin InGaAs interface layer of less than
100 Å is inserted. At the InGaAs to GaAsSb interface, TEGa was switched to the run line after one second of TBAs and TMSb purging. Detailed studies on the interface layer will be reported elsewhere. With a nominal V/III ratio of 2, the Sb distribution coefficient for undoped GaAsSb layers lattice matched to InP is approximately equal to 0.9. This facilitates composition control for GaAs12xSbx . As the V/III ratio increases, the distribution coefficient decreases.5,6 Under these growth conditions, mirror-like surfaces were obtained for GaAs0.5Sb0.5 grown on InP substrates. A typical atomic force microscopy ~AFM! image of a GaAs0.5Sb0.5 surface is shown in Fig. 1~a! after 700 Å of growth. The root mean square ~RMS! surface roughness is less than 5 Å, but the surface is clearly textured. The reason for the unusual GaAsSb morphology is not clear yet. After the overgrowth of 1000 Å InP on the GaAsSb layer, the surface texture disappears and atomic steps are clearly observed, as shown in Fig. 1~b!. Figure 2 shows the ~004! High resolution x-ray diffraction ~XRD! curve for a 700 Å GaAsSb layer grown on InP: no signs of phase separation can be detected. The XRD peak width is comparable to that simulated by the dynamical x-ray diffraction theory. This clearly shows that high-quality GaAsSb can be grown although a large miscibility gap is predicted at this composition.
By Edgar Alberto Servita 18.856.338
CAF
HIGH-ELECTRON-MOBILITY
HIGH-ELECTRON-MOBILITY transistors (HEMTs) and heterojunction bipolar transistors (HBTs) have attracted many attentions in high speed and power applications due to the superior transport properties. As compared to AlGaAs pseudomorphic HEMTs (PHEMTs), InGaP-related devices with the following advantages, such as higher band gaps, higher valence-band discontinuity [1], negligible deep-complex (DX) centers [2], excellent etching selectivity between InGaP and GaAs, good thermal stabilities, higher Schottky barrier heights [3], and so on. Particularly, the use of an undoped InGaP insulator takes the advantages of its low DX centers and low reactivity with oxygen [4], [5], which may still suffer from a high gate leakage issue. In order to inhibit the gate leakage issue, increase the power handling capabilities, and improve the breakdown voltages, a MOS structure has been widely investigated. However, there is still a lack of reliable native oxide films growing on InGaP, and very few papers have reported on InGaP/InGaAs MOS-PHEMTs. Over the past few years, an alternative technique named liquid-phase chemical-enhanced oxidation (LPCEO) to growreliable native oxide films on GaAs [6], [7], Si [7], InP [7],
AlGaAs [8], and InGaAs [9] has been reported. This is an easy, low-cost, and low-temperature (30 C–70 C) technique to grow uniform and smooth native oxide films on GaAs-based materials. Moreover, in the liquid-phase oxidation system, neither anodic equipment nor an assisting energy source is needed. According to the preliminary studies of this technique, some issues were addressed [10], [11]. In this work, a thin InGaP native oxide layer prepared by the liquid-phase oxidation as the gate dielectric for InGaP/InGaAs MOS-PHEMT application is demonstrated.
EXPERIMENTAL
The details of the oxidation system were reported earlier in [6] and [7]. Although liquid-phase oxidation on InGaP material has a much slower oxidation rate, less than 10 nm/h, compared with that of the GaAs material, it is still feasible to grow a thin oxide film. The oxidation rate becomes significantly saturated when the oxidation time is longer than an hour, which is measured using a Veeco Instrument DEKTAK and confirmed by scanning electron microscopy (SEM). The oxide film is mostly composed of InPO -like and Ga oxide, which is confirmed by the values of the peak energy and energy separations of X-ray photoelectron spectroscopy (XPS) between main core levels schematically shows the PHEMT structure grown by the metal–organic chemical vapor deposition (MOCVD) on a semi-insulating GaAs substrate. Hall measurement indicates that the electron mobility is 4000 cm /V s and the electron sheet density is cm at room temperature. The device isolation was accomplished by mesa wet etching down to the buffer layer. Ohmic contacts of the Au-Ge-Ni metal were deposited by evaporation and then patterned by lift-off processes, followed by rapid thermal annealing. The depth of gate recess is 110 nm for the reference PHEMT and 100 nm for the MOS-PHEMT. After etching the capping layer and partial Schottky layer, a LPCEO growth solution was used to generate the gate oxide for the MOS-PHEMT at 50 C for 30 min. Finally, the gate electrode was formed with Au. Moreover, the oxide, as illustrated in the figure, also selectively and simultaneously passivated the isolated surface sidewall.
By Edgar Alberto Servita 18.856.338
CAF
Metal-Semiconductor Field Effect Transistor (MESFETs)
Metal-Semiconductor Field Effect Transistor (MESFETs)
The Metal-Semiconductor-Field-Effect-Transistor (MESFET) consists of a conducting channel
positioned between a source and drain contact region as shown in the Figure 3.6.1. The carrier
flow from source to drain is controlled by a Schottky metal gate. The control of the channel is
obtained by varying the depletion layer width underneath the metal contact which modulates the
thickness of the conducting channel and thereby the current between source and drain.
Structure of a MESFET with gate length, L, and channel thickness, d.
The key advantage of the MESFET is the higher mobility of the carriers in the channel as
compared to the MOSFET. Since the carriers located in the inversion layer of a MOSFET have a
wavefunction, which extends into the oxide, their mobility - also referred to as surface mobility -
is less than half of the mobility of bulk material. As the depletion region separates the carriers
from the surface their mobility is close to that of bulk material. The higher mobility leads to a
higher current, transconductance and transit frequency of the device.
The disadvantage of the MESFET structure is the presence of the Schottky metal gate. It limits
the forward bias voltage on the gate to the turn-on voltage of the Schottky diode. This turn-on
voltage is typically 0.7 V for GaAs Schottky diodes. The threshold voltage therefore must be
lower than this turn-on voltage. As a result it is more difficult to fabricate circuits containing a
large number of enhancement-mode MESFET.
The higher transit frequency of the MESFET makes it particularly of interest for microwave
circuits. While the advantage of the MESFET provides a superior microwave amplifier or circuit,
the limitation by the diode turn-on is easily tolerated. Typically depletion-mode devices are used
since they provide a larger current and larger transconductance and the circuits contain only a
few transistors, so that threshold control is not a limiting factor. The buried channel also yields a
better noise performance as trapping and release of carriers into and from surface states and
defects is eliminated.
The use of GaAs rather than silicon MESFETs provides two more significant advantages: first,
the electron mobility at room temperature is more than 5 times larger, while the peak electron
velocity is about twice that of silicon. Second, it is possible to fabricate semi-insulating (SI)
GaAs substrates, which eliminates the problem of absorbing microwave power in the substrate
due to free carrier absorption.
By Edgar Alberto Servita 18.856.338
CAF
RF and Microwave Power Amplifier and Transmitter Technologies
RF and microwave power amplifiers and transmitters are used in a wide variety of applications including wireless communication, jamming, imaging, radar, and RF heating. This article provides an introduction and historical background for the subject, and begins the technical discussion with material on signals, linearity, efficiency, and RF-power devices. At the end, there is a convenient summary of the acronyms used—this will be provided with all four installments. Author affiliations and contact information are also provided at the end of each part.
The generation of significant power at RF and microwave frequencies is required not only in wireless communications, but also in applications such as jamming, imaging, RF heating, and miniature DC/DC converters. Each application has its own unique requirements for frequency, bandwidth, load, power, efficiency, linearity, and cost. RF power can be generated by a wide variety of techniques using a wide variety of devices. The basic techniques for RF power amplification via classes A, B, C, D, E, and F are reviewed and illustrated by examples from HF through Ka band. Power amplifiers can be combined into transmitters in a similarly wide variety of architectures, including linear, Kahn, envelope tracking, outphasing, and Doherty.
Linearity can be improved through techniques such as feedback, feedforward, and predistortion. Also discussed are some recent developments that may find use in the near future.
A power amplifier (PA) is a circuit for converting DC input power into a significant amount of RF/microwave output power. In most cases, a PA is not just a small-signal amplifier driven into saturation. There exists a great variety of different power amplifiers, and most employ techniques beyond simple linear amplification.
A transmitter contains one or more power amplifiers, as well as ancillary circuits such as signal generators, frequency converters, modulators, signal processors, linearizers, and power supplies. The classic architecture employs progressively larger PAs to boost a low-level signal to the desired output power.
However, a wide variety of different architectures in essence disassemble and then reassemble the signal to permit amplification with higher efficiency and linearity.
Modern applications are highly varied.
Frequencies from VLF through millimeter wave are used for communication, navigation, and broadcasting. Output powers vary from 10 mW in short-range unlicensed wireless systems to 1 MW in long-range broadcast transmitters.
Almost every conceivable type of modulation is being used in one system or another. PAs and transmitters also find use in systems such as radar, RF heating, plasmas, laser drivers, magnetic-resonance imaging, and miniature DC/DC converters.
HISTORICAL DEVELOPMENT
Spark, Arc, and Alternator
I
n the early days of wireless communication (from 1895 to the mid
1920s), RF power was generated by spark, arc, and alternator techniques.
The original RF-power device, the spark gap, charges a capacitor to a high voltage, usually from the AC mains. A discharge (spark) through the gap then rings the capacitor, tuning inductor, and antenna, causing radiation of a damped sinusoid.
Spark-gap transmitters were relatively inexpensive and capable of generating 500 W to 5 kW from LF to MF [1].
The arc transmitter, largely attributed to Poulsen, was a contemporary of the spark transmitter. The arc exhibits a negative-resistance characteristic which allows it to operate as a CW oscillator (with some fuzziness). The arc is actually extinguished and reignited once per RF cycle, aided by a magnetic field and hydrogen ions from alcohol dripped into the arc chamber. Arc transmitters were capable of generating as much as 1 MW at LF [2].
The alternator is basically an AC generator with a large number of poles. Early RF alternators by Tesla and Fessenden were capable of operation at LF, and a technique developed by Alexanderson extended the operation to LF [3]. The frequency was controlled by adjusting the rotation speed and up to 200 kW could begenerated by a single alternator. One
such transmitter (SAQ) remains operable at Grimeton, Sweden.
Vacuum Tubes
With the advent of the DeForest audion in 1907, the thermoionic vacuum tube offered a means of electronically generating and controlling RF signals. Tubes such as the RCA UV-204 (1920) allowed the transmission of pure CW signals and facilitated the transition to higher frequencies of operation.
Younger readers may find it convenient to think of a vacuum tube as a glass-encapsulated high-voltage FET with heater. Many of the concepts for modern electronics, including class-A, -B, and -C power amplifiers, originated early in the vacuumtube era. PAs of this era were characterized by operation from high voltages into high-impedance loads and by tuned output networks. The basic circuits remained relatively unchanged throughout most of the era.
Vacuum tube transmitters were dominant from the late 1920s through the mid 1970s. They remain in use today in some high power applications, where they offer a relatively inexpensive and rugged means of generating 10 kW or more of RF power.
Discrete Transistors
Discrete solid state RF-power devices began to appear at the end of the 1960s with the introduction of silicon bipolar transistors such as the 2N6093 (75 W HF SSB) by RCA. Power MOSFETs for HF and VHF appeared in 1974 with the VMP-4 by Siliconix. GaAs MESFETs introduced in the late 1970s offered solid state power at the lower microwave frequencies.
The introduction of solid-state RF-power devices brought the use of lower voltages, higher currents, and relatively low load resistances. Ferrite-loaded transmission line transformers enabled HF and VHFPAs to operate over two decades of bandwidth without tuning. Because solid-state devices are temperaturesensitive, bias stabilization circuits were developed for linear PAs. It also became possible to implement a variety of feedback and control techniques through the variety of opamps and ICs.
Solid-state RF-power devices were offered in packaged or chip form. A single package might include a number of small devices. Power outputs as high as 600 W were available from a single packaged push-pull device (MRF157). The designer basically selected the packaged device that best fit the requirements. How the transistors were made was regarded as a bit of sorcery that occurred in the semiconductor houses and was not a great concern to the ordinary circuit designer.
Custom/Integrated Transistors
The late 1980s and 1990s saw a proliferation variety of new solidstate devices including HEMT, pHEMT, HFET, and HBT, using a variety of new materials such as InP, SiC, and GaN, and offering amplification at frequencies to 100 GHz or more. Many such devices can be operated only from relatively low voltages. However, many current applications need only relatively low power. The combination of digital signal processing and microprocessor control allows widespread use of complicated feedback and predistortion techniques to improve efficiency and linearity. Many of the newer RF-power devices are available only on a madeto- order basis. Basically, the designer selects a semiconductor process and then specifies the size (e.g., gate periphery). This facilitates tailoring the device to a specific power level, as well as incorporating it into an RFIC or MMIC.
By Edgar Alberto Servita 18.856.338
CAF
Monolithic Microwave Integrated Circuits
Monolithic Microwave Integrated Circuits
A. General Description
Monolithic Microwave Integrated Circuits (MMICs) are used in satellite systems
that require smaller, less expensive circuits or when the parasitic reactance inherent in
hybrid integrated circuits degrades the circuit performance, typically in the upper
microwave and the millimeter-wave spectrum. Examples of systems that use MMICs are
receivers and transmitters for communications, phased-array antennas where small size
and uniform circuit performance are required, and sensors and radars that operate at high
frequencies. The types of circuits required for each of these systems are illustrated by
examining the simple receiver and transmitter systems shown in Figures 3-38 and 3-39,
respectively. In both schematics, a phase shifter—which may be placed in either the
local oscillator (LO), the RF, or the IF portion of the system—has been added to make
the system perform as if each circuit were coupled to a single radiating element of a
phased-array antenna. For non-phased-array applications, the schematic is unchanged
except for the removal of the phase shifter. A photograph of a completely monolithic 30-
GHz receiver is shown in Figure 3-40. Although the high level of circuit integration
illustrated in Figure 3-40 decreases the packaging and interconnect costs, this integration
is not necessary or common. Instead, each function of the system is typically fabricated
on an individual die to permit the optimization of the material system and device type for
each application. Regardless of the level of circuit interconnection, the reliability of the
system is dependent on the continuous operation of each circuit.
Research Center.)
This is understood by examining the receiver circuit shown in Figure 3-38. The
input (RF) signal typically has a very low power level that may be close to the noise
floor. The low-noise amplifier (LNA) amplifies the received signal while at the same
time introduces very little new noise. If the gain of the LNA is sufficiently large, the
noise contributions of the rest of the system will be small since the noise created by later
circuits is divided by the gain of the LNA. Thus, the LNA gain and noise figure, the
measure of noise added by the LNA, determine the receiver noise characteristics. If the
receiver has poor noise characteristics, it will not be able to receive weak signals. The
signal may then pass through a narrow-band filter and into the mixer. The LO generates
a signal that is also fed into the mixer. The mixer combines the two signals through a
nonlinear device, such as a MESFET or diode, and generates a signal at the intermediate
frequency (IF) of fRF – fLO or fLO – fRF and harmonics of the IF, RF, and LO frequencies.
All but the desired IF components must be filtered out. The conversion efficiency of the
mixer is usually dependent on the LO drive power. In addition, a variation in the LO
frequency will cause a shift in the IF that may cause the signal to be attenuated in the
narrow-band filters that are part of the mixer. If the system is to be associated with a
phased-array antenna, the direction and shape of the main beam radiated or received by
the antenna is dependent on the relative phase shift and power level of each transmitter
(and receiver). The relative phase of each radiating element is set by the phase shifter.
Thus, if the phase shift through the circuit varies because of unexpected conditions, the
efficiency of the entire antenna will degrade. It is thus seen that a parametric shift by any
of the components may cause the entire system to fail.
The phase shifter, local oscillator, and mixer circuits are common to the receiver
and transmitter with the exception of a shift in the design frequency. The real difference
Amplifiers
Both low-noise and power amplifiers are used to increase the power of the RF
signal. In almost all systems, this is accomplished by using the transconductance of
MESFETs and HEMTs or the current gain of HBTs. The amount of signal increase is
called "gain" and is usually given in dB, where gain in dB = 10 log (gain). For example,
if the output power is twice the input power, the amplifier has 3 dB of gain. Typically,
the input power and the output power are also specified in dB, permitting the output
power to equal the sum of the input power and the gain. This ideal operation of an
amplifier is accurate for low power levels. Unfortunately, as power levels increase, the
amplifier becomes nonlinear. In the nonlinear region of operation, the output power is
less than the sum of the input power and the amplifier gain in the linear region, or it can
be stated that the amplifier gain is lower in the nonlinear region. Figure 3-41 shows a
typical amplifier characteristic. The point at which the output power drops by 1 dB from
the linearly extrapolated value is called the 1-dB compression point [1]. This value
separates small-signal or linear amplifiers from large-signal or power amplifiers. Note
that this is also the criteria used to differentiate small-signal and large-signal transistors,
since a transistor can be viewed as a simple, unmatched amplifier. This differentiation is
important in determining the failure mechanisms that need to be addressed and the type
of reliability tests that should be performed.
Power Amplifiers
Power amplifiers, by their very nature, must handle high input and output powers.
The maximum voltage swing of the input signal is limited by the breakdown voltage of
the transistor, and thus transistors with high breakdown voltages are required. The
current through each transistor is limited by the resistance in the gate or emitter of FETs
and HBTs, respectively, since ohmic losses are converted to heat, which decreases the
device's reliability. To increase the current handling capability of the device, power
transistors combine many gates or emitters in parallel. This parallel combination
increases the total gate width or emitter area and decreases the resistance, while at the
same time increases the difficulty in matching the input impedance of the transistor to the
output impedance of the prior stage. In addition, the spacing required between the
transistor elements to permit sufficient thermal dissipation creates large devices that are
more difficult to maintain with a uniform voltage [3]. To dissipate the heat from the
transistors, power amplifiers are fabricated on thin wafers, less than 100 mm thick and
typically between 25 and 50 mm, to reduce the thermal path between the transistor's
active region and a good heat sink, such as a metal or diamond carrier. Generally,
thermal constraints limit the design and performance of power amplifiers more than
frequency constraints. Thus, the efficiency of power amplifiers is one of the most critical
specifications, especially in space applications where satellite power is limited, where
dissipation of the thermal load requires heat sinks that increase the system weight, and
where circuit heating can decrease reliability.
Low-Noise Amplifiers\
Since low-noise amplifiers are used on the front end of receivers, they are
designed to handle very low power levels. Thus, the thermal problems and high bias
currents and voltages that affect power amplifier reliability are generally not a concern
for LNA designers. The most important criterion in specifying or measuring an LNA's
performance is the noise figure, and since HEMTs and PHEMTs have the lowest noise
figure, they are used in almost all LNAs. To minimize the noise figure, small gate
lengths and low parasitic gate and source resistances are required [4]. Thus, state-of-theart
LNAs are usually comprised of 0.1 to 0.25 mm gate-length HEMTs or PHEMTs, and
the reliability concerns—such as gate metal sinking and ohmic contact diffusion (see
Chapter 4)—arising from small gate lengths and corresponding small channel thicknesses
are the most important.
To decrease the noise figure of the system, it is important to reduce the circuit
losses, especially before the first stage of the LNA. This includes the package feed losses
and transmission line losses from the antenna since they introduce noise into the system
before the LNA. Besides reducing the circuit losses, noise can be reduced by operating
the amplifier at lower temperatures and lower bias currents and voltages. Lastly, the
noise figure of the LNA is dependent on the matching circuits, which are designed with
an input matching network that minimizes the noise figure and an output matching
network to maximize the gain. The optimum input matching network can be found
through noise parameter measurements of the HEMT. From these measurements, an
equivalent circuit model of the HEMT that includes noise sources can be generated.
Mixers
Mixers convert an input signal at one frequency to an output signal at another
frequency to permit filtering, phase shifting, or some other data processing operation at a
frequency more easily implemented by the circuits. For example, a system may require
the data to be received at W-band, 75 to 110 GHz, but W-band filters have a low Q or a
high loss, which degrades the receiver noise characteristics. Therefore, it may be
advantageous to shift the received signal's frequency to a lower value where low-loss
filters are possible. Ideally, this operation is accomplished without degrading the input
signal's amplitude or introducing additional noise.
Oscillators
Oscillators generate microwave energy for communications, radars, and
navigation systems. For example, modulators, superheterodyne receivers, and phasedlocked
loops depend on a good microwave source to function. In principal, any amplifier
could be made into an oscillator by providing positive feedback to the input terminals so
that the reflection coefficient of the amplifier is greater than one. More often than not,
this is accidentally done by amplifier designers. Therefore, an oscillator is basically an
LNA with a feedback loop that introduces delay-of-integer multiples of 2p. The choice
of the load and terminating impedance to achieve this condition should also guarantee the
proper oscillation frequency and maximize the efficiency or RF power delivered to a
load. In general, there are two types of oscillators: fixed-frequency oscillators designed
to operate at a single frequency and variable-frequency oscillators or voltage-controlled
oscillators (VCOs) with tuning circuits that change the oscillation frequency. The
schematic of a simple oscillator is shown in Figure 3-45. It consists of a transistor with
feedback between the gate and drain, an output matching circuit, and a resonant structure
on the input. Oscillator performance specifications or figures of merit that affect the
system reliability include phase noise and thermal stability
Phase Shifters
Phase shifters are used to impart a repeatable and controllable change of phase to
a microwave signal with no effect on the signal's amplitude. Although they are usually
associated with phased-array antennas, where they are used to control the beam shape and
direction, they are also used in communication systems, radar systems, and microwave
instrumentation. Two methods are commonly used to change the phase in MMICs. The
first method switches the signal between a short and a long length of transmission line to
develop a phase shift of b where b is the propagation constant of the transmission line
and is the differential transmission line length. This type of phase shifter is called a
switched-line phase shifter and is a true time-delay phase shift. The second method
changes the reactance of a transmission line, which changes the propagation constant
along the line. The implementation of MMIC phase shifters is broadly characterized as
either reflection type or transmission type.
By Edgar Alberto Servita 18.856.338
CAF
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